Superconducting multi-bit digital mixer

ABSTRACT

A superconducting multi-bit digital mixer, designed using rapid single flux quantum (RSFQ) logic, for multiplying two independent digital streams, at least one of these comprising a plurality of parallel bit lines, wherein the output is also a similar plurality of bit lines. In a preferred embodiment, one of the digital streams represents a local oscillator signal, and the other digital stream digital radio frequency input from an analog-to-digital converter. The multi-bit mixer comprises an array of bit-slices, with the local oscillator signal generated using shift registers. This multi-bit mixer is suitable for an integrated circuit with application to a broadband digital radio frequency receiver, a digital correlation receiver, or a digital radio frequency transmitter. A synchronous pulse distribution network is used to ensure proper operation at data rates of 20 GHz or above.

RELATED APPLICATION

The present application claims the benefit of priority of ProvisionalPatent Application 61/369,927, “Superconducting andsemiconductor-superconductor hybrid systems and devices, and methods forpackaging and manufacturing thereof”, filed Aug. 2, 2010, the entiretyof which is expressly incorporated herein by reference.

STATEMENT OF GOVERNMENT INTEREST

This invention was developed in part under U.S. Office of Naval ResearchContracts N00173-07-C-4002 and N00014-08-C-0603, and the U.S. Governmenthas certain rights therein.

FIELD OF THE INVENTION

The present invention relates to digital radio-frequency receivertechnology, and more particularly a digital downconverter circuitimplemented using superconducting electronic components.

BACKGROUND OF THE INVENTION

FIGS. 1A and 1B show a classic analog radio receiver and a fullydigital, radio frequency receiver. The core of a conventional radioreceiver comprises a circuit that mixes the input radio signal from theantenna with a local oscillator (LO). Both the LO and the mixer areanalog devices. The mixer generates frequency components correspondingto the sum and difference of the input frequencies of the radiofrequency (RF) signal and the LO. An analog low-pass filter (LPF) isused to isolate the down-converted baseband signal, which can then bedigitized in a baseband analog-to-digital converter (ADC). FIG. 1A showsa quadrature receiver (also called an I&Q receiver) with a localoscillator having in-phase and quadrature (90-degree phase shifted)outputs, which are mixed with the input RF signal to produce I and Qsignals, respectively. The baseband I and Q signals can be combined in aDigital Baseband Processor to demodulate one or more communicationsignals of interest.

In a digital radio frequency receiver (sometimes known as a softwareradio or software-defined radio or SDR), the conversion to digital iscarried out directly on the RF signal coming from the antenna. Asindicated in FIG. 1B, the same set of operations need to be carried outas in FIG. 1A, but now in the digital domain. These include a digitalLO, a digital mixer (really a digital multiplier), and a digitaldecimation filter (DDF, the digital equivalent of a low-pass filter).These digital operations are implemented without noise, interference, ornon-ideal behavior, and may be easily reprogrammed for differentcommunication parameters. Furthermore, the digital radio frequencyreceiver is compatible with a very broad RF bandwidth, which maycomprise a plurality of communication signals. These advantages areincreasingly desirable for modern RF receivers. However, a requirementof the digital radio frequency approach is that for optimum performance,the RF ADC must operate at a very high sample rate (typicallymulti-GHz), and in some cases also needs a plurality of parallel databits to increase the dynamic range. This high rate requirement isbecause the sampling should be above the Nyquist rate (twice the highestfrequency component of the signal), but in some cases the radiofrequency signal can be undersampled, so long as the sampling isprecisely timed. Assuming that undersampling is not, or cannot be used,the following mixer and digital filter must also operate at the veryhigh data rate, with a plurality of data bits. These requirements havelimited the availability of such a digital radio frequency receiver.See, e.g., U.S. Pat. Nos. 7,956,640; 7,928,875; 7,903,456; 7,876,869;7,728,748; 7,701,286; 7,680,474; 7,598,897; 7,443,719; 7,362,125;7,313,199; 7,280,623; 6,781,435; Muhammad, K.; Staszewski, R. B.;Leipold, D.; “Digital RF processing: toward low-cost reconfigurableradios”, IEEE Communications Magazine, August 2005, Volume: 43 Issue: 8,pages: 105-113; Brock, D. K.; Mukhanov, O. A.; Rosa, J.; “Superconductordigital RF development for software radio”, IEEE CommunicationsMagazine, February 2001, Volume: 39 Issue: 2, page(s): 174-179;Saulnier, G. J.; Puckette, C. M., IV; Gaus, R. C., Jr.; Dunki-Jacobs, R.J.; Thiel, T. E.; “A VLSI demodulator for digital RF networkapplications: theory and results”; IEEE Journal on Selected Areas inCommunications, October 1990, Volume: 8 Issue: 8, page(s): 1500-1511,each of which is expressly incorporated herein by reference.

Superconducting electronics using rapid-single-flux-quantum (RSFQ)circuits provide the world's fastest digital circuits (operating atrates of up to 40 GHz and higher). And indeed, very fast RSFQ ADCs,digital mixers, and digital filters have been demonstrated. However,most of these circuits have been relatively small circuits, operating ona single bitstream of data. See, for example, U.S. application Ser. No.11/966,897, “Oversampling Digital Receiver for Radio Frequency Signals”,D. Gupta, and U.S. Pat. No. 7,280,623, “Digital RF Correlator forMultipurpose Digital Signal Processing”, D. Gupta, et al., both of whichare expressly incorporated herein by reference.

For the case of a superconducting digital mixer, several designs of afast single-bit RSFQ digital mixer were presented by Kirichenko et al.(“Superconducting Digital Mixer”, U.S. Pat. No. 7,680,474), expresslyincorporated herein by reference, one of which is shown schematically inFIG. 2. This is based on the mapping convention of FIG. 3A, where binarymultiplication of a bipolar signal maps onto the exclusive OR (XOR)function of a unipolar binary signal.

To improve this technology and to make it more practical, multibitimplementations must be developed. The present invention focuses on thedevelopment of an integrated multibit digital mixer based on RSFQtechnology, building upon the foundations of an earlier single-bit mixerof Kirichenko.

SUMMARY OF THE INVENTION

The present technology provides an XOR-based mixer cell similar to thatin FIG. 2, which can be used to design, and lay out as an integratedcircuit, a multi-bit mixer cell which can also perform at data rates oftens of GHz. Two preferred embodiments are presented; an n×1 mixer inwhich an n-bit digitized data stream is mixed with a 1-bit LO; and a 1×kmixer in which a 1-bit data stream is mixed with an k-bit LO, where nand k represent integers greater than 1. Both embodiments are organizedinto single-bit-slices that can be tiled together in an IC to scale to astructure with an arbitrary number of bits. Furthermore, bothembodiments make use of a shift register (SR) as a code generator forthe LO, incorporate a synchronous pulse distribution network (SPDN) forproper multibit timing, and are otherwise very similar in design andperformance. The data and the LO may operate at different clock rates,since the mixer elements themselves are asynchronous. Furthermore, it isshown how the multibit mixer may be integrated with a multi-bit ADC anda multi-bit DDF to form an improved multi-bit digital radio frequencyreceiver.

Note that in these cases, each bit is processed separately, in parallel;no carry operations are required for this procedure. This is illustratedwith the 3-bit multiplication table in FIG. 3B, using the same mappingconvention per bit as in FIG. 3A. Note also that this mapping cannot bedirectly extended to full n×k multiplication, where carry operationswould be necessary, making the required circuit much more complicated.

A block diagram for an I&Q digital 1×k mixer according to the inventionis shown in FIG. 4, with two 1×k mixers, data and LO clock lines. Eachmixer is comprised of k slices (typically arrayed vertically), eachslice comprised of a cell structure such as that shown in FIG. 5. Thisgenerates a k-bit mixer output for each of the two (I and Q) channels.Unlike the block diagram of FIG. 1B, which shows a separate LOgenerator, the circuit of FIGS. 4 and 5 preferably integrates themulti-bit LO generator elements for each bit into the mixer sliceitself. Note specifically the shift register SR on the right of FIG. 5,which comprises the LO generator for the given bit. Prototype mixercircuits with a 1×3 (3-bit LO) and a 3×1 (3-bit input data) arepresented below in more detail, with IC fabrication and experimentaltest results confirming proper operation.

An alternative embodiment of a 1×k multi-bit mixer is also presented,which comprises an external LO generator, rather than the internal LOgenerator as shown in FIGS. 4 and 5. In this example, with a blockdiagram as shown in FIG. 6, the LO multi-bit code is synthesized inadvance and stored in a room-temperature memory with a fast readout.This multi-bit code is then serialized and sent out to thesuperconducting circuit as a single-bit sequence (in order to avoidtiming skew between the bits). This LO code is then deserialized andresynchronized in a fast cache memory that is integrated with themulti-bit mixer. Further details of this embodiment are described below.

Multi-bit digital mixer circuits of the invention are not limited toapplication to a digital downconverter in a digital radio frequencyreceiver. They may also be used in a digital correlation receiver, wherea more general synthesized digital signal may be used instead of adigital local oscillator. Alternatively, a similar multi-bit digitalmixer may be used as the digital upconverting element in a digital radiofrequency transmission system.

It is therefore an object to provide a multi-bit digital mixercomprising at least one Josephson junction, configured in a circuit toreceive a first digital input signal and a second digital input signaland to generate a multi-bit parallel output signal representing themultiplication product of the first digital input signal and the seconddigital input signal.

The second digital input signal may be received from a digital localoscillator. The digital LO signal may be generated using a shiftregister.

The first digital input signal may be communicated over at least onematched pair of complementary binary inputs.

The circuit may comprise a plurality of bit slices, having acorresponding bit slice for each parallel bit of the output signal.

The circuit may comprise at least one respective XOR-based mixer cellfor each bit of the multi-bit parallel output signal.

Each of the first digital input signal and the second digital inputsignal, or both, may comprise a multi-bit signal. The multi-bit signalsmay be parallel or serial.

At least one of the first and second digital input signals may comprisea multi-bit parallel signal which incorporates a synchronous pulsedistribution network (SPDN) for proper multibit timing. The SPDN mayprovide global bit synchronization for both input signals.

The circuit may comprise at least one asynchronous RSFQ mixer. Forexample, the mixer may be an asynchronous XOR device.

The first digital input signal and the second digital input signal mayhave respectively different clock rates.

The first digital input signal may receive a signal derived from ananalog to digital converter, and the second digital input signalreceives a signal derived from a digital local oscillator. The analog todigital converter may receive a modulated radio frequency signal, andthe circuit may be configured to mix the modulated radio frequencysignal to generate the multi-bit parallel output signal representing atleast a difference frequency of the modulated radio frequency signal andthe digital local oscillator.

The circuit may be configured to generate the multi-bit parallel outputsignal representing the multiplication product of the first digitalinput signal and the second digital input signal substantially without acarry operation.

The circuit may further comprise a semiconductor circuit, configured togenerate at least one of the first digital input signal and the seconddigital input signal, further comprising an interface circuit configuredto present information from the semiconductor circuit to the circuitcomprising the at least one Josephson junction.

The circuit may also comprise a semiconductor circuit, furthercomprising an interface circuit configured to present information fromthe circuit comprising at least one Josephson junction to thesemiconductor circuit.

The circuit may be configured as a digital down converter of a digitalradio frequency receiver receiving analog radio frequency informationmodulated on a carrier having a carrier frequency of at least 250 MHz,comprising a superconducting analog to digital converter having asampling rate of at least 1 GHz and a multi-bit output. Thus, forexample, a 40 GHz analog to digital converter may be processed toproduce a 3-bit signal at a rate of 5 GHz.

The circuit may be configured to produce an updated multi-bit paralleloutput signal at a rate of at least 20 GHz.

The circuit may be configured as a receiver for receiving quadraturephase modulated signals, further comprising an analog to digitalconverter digitizing a radio frequency signal at a digital sample rateof at least 1.5 GHz, wherein at least two local oscillator signals areprovided by at least one shift register configured as a code generatorto each of at least two multi-bit digital mixers, each multi-bit mixerbeing configured to produce a multi-bit parallel output signalrepresenting the multiplication product of the digital output of theanalog to digital converter and a respective representation of the localoscillator signals, wherein the local oscillator signals arerespectively time-shifted replicas of each other.

Another object provides a digital mixing method, comprising: receivingtwo asynchronous rapid single quantum flux (RSFQ) signals, at least oneof the received RSFQ signals comprising a plurality of parallel bits,wherein each signal bit is represented as complementary signal pairs;ensuring proper timing of the plurality of parallel bits of the twoasynchronous RSFQ signals with a synchronous pulse distribution network(SPDN); and mixing the properly timed two asynchronous RSFQ signals withan array of XOR-based asynchronous mixer cells, at least one for eachparallel bit line.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1A shows a block diagram of a conventional I&Q radio receiver withan analog front end, comprising an analog local oscillator (LO) and ananalog mixer.

FIG. 1B shows a block diagram of a digital radio frequency receiver (I&Qchannels) with a broadband ADC digitizing the RF signal directly,comprising a digital LO and a digital mixer.

FIG. 2 shows the cell diagram of a 1-bit asynchronous digital mixer ofthe prior art, according to Kirichenko et al. (U.S. Pat. No. 7,680,474),based on a streaming-XOR functionality

FIG. 3A shows the logic table for a 1-bit XOR-based digital mixer,mapping bipolar binary multiplication to an XOR function of unipolarsignals.

FIG. 3B shows the logic table for a 1×3 bit XOR based digital mixer,mapping bipolar binary multiplication to bit-parallel XOR functions.

FIG. 4 shows the block diagram of an I&Q digital mixer for a 1-bit datasignal with a k-bit LO (1×k mixer).

FIG. 5 shows the cell diagram of a bit-slice of a 1×k digital mixer withintegrated LO of FIG. 4.

FIG. 6 shows the block diagram of an alternative digital radio frequencyreceiver with a superconducting multi-bit mixer and an externallygenerated LO.

FIG. 7 shows the block diagram of an improved 1×1 bit I&Q mixer,modified from that in FIG. 2.

FIG. 8 shows the cell diagram of a shift register-based LO forintegration with a bit-slice of the 1×k digital mixer of FIG. 5.

FIG. 9 shows the block diagram of an I&Q mixer for an n-bit data signalwith a 1-bit LO (n×1 mixer).

FIG. 10 shows the cell diagram of a bit slice of an n×1 digital mixer ofFIG. 9.

FIG. 11 shows the chip layout of a prototype 1×3 I&Q mixer test chip.

FIG. 12 shows the LO code and data used to test the 3×1 mixer (A) andthe 1×3 mixer (B).

FIG. 13 shows plots of the LO magnitude vs. time, for a 3×1 mixer (A)and a 1×3 mixer (B).

FIG. 14 shows the results of experimental testing of the 1×3 mixer chip,confirming proper operation.

FIG. 15 shows the results of experimental testing of the 3×1 mixer chip,confirming proper operation.

FIG. 16 shows the layout of a lowpass all-digital receiver chipincorporating a 1×3 I&Q mixer circuit.

FIG. 17 shows the block diagram of an I&Q digital mixer for a 1-bit datasignal with a k-bit LO (1×k mixer), similar to FIG. 4, but with a cacheblock (comprising deserializers DS) receiving an externally generated LOcode as shown in FIG. 6.

FIG. 18 shows the cell diagram of a bit-slice of a 1×k digital mixerwithout integrated LO of FIG. 17.

FIG. 19 shows the block diagram of an RSFQ cache memory circuit formulti-bit LO re-synchronization, as in FIGS. 6 and 17.

FIG. 20 shows a portion of the chip layout for a 1×3 digital mixer withexternal LO generator and on-chip integrated cache memory.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Digital down-conversion requires multiplication of a digitized radiofrequency signal with a digital local oscillator signal. Afirst-generation digital radio frequency receiver (following FIG. 1B)was developed by HYPRES, Inc. (see D. Gupta, et al., “DigitalChannelizing Radio Frequency Receiver,” IEEE Trans. Appl. Supercond.,vol. 17, no. 2, pp. 430-437, June 2007, expressly incorporated herein byreference), also known as an All-Digital Receiver (ADR)). This includeda 1×1-bit mixer to process a 1-bit data stream from a modulator (ADC)and a 1-bit square wave from a local oscillator. The next steps in thedevelopment of improved digital radio frequency receivers are theprocessing of either n-bit data streams from advanced low pass (LP) orband pass (BP) modulators, or the use of a k-bit local-oscillatorstream.

The present approach is based on the design and successful testing ofthe 1×1-bit mixer (A. Kirichenko et al., U.S. Pat. No. 7,680,474) shownin FIG. 2. The multiplication table for the 1×1-bit mixer and itsmapping convention for a bipolar data stream (D) and a particular localoscillator code (C) are shown in FIG. 3A. This allows one to treatmultiplication as an exclusive-OR (XOR) function with data and codeinputs.

Note that the original design of a 1×1-bit mixer (FIG. 2) differsslightly from the improved one shown in FIG. 7. In its original version,the D-cell (D flip flop or DFF) and inverter (N cell) were combined intoa D-cell with complementary outputs (DFFC), and a multiplexer cell (MUX)was used instead of a matched pair of RSN cells (RS flip-flops withnondestructive readout). Functionally these two designs are largelyequivalent, and while the original design was slightly more compact, itwas less robust and more susceptible to stray magnetic fields, limitingits performance margins. Furthermore, the logic for generating the fourphases of the local oscillator was modified using the Multiphase ClockCircuit of D. Kirichenko (U.S. Pat. No. 7,786,786, expresslyincorporated herein by reference) to overcome the susceptibility of theearlier design to any error in the toggle flip-flop (TFF, or simply ‘T’)tree, causing a permanent phase slip between I and Q streams. Thismultiphase clock circuit comprised a chain of D-flip-flops.

The rule of action for the design in FIG. 7 can also be formulated as:data bit D reads complement code C, complement data D reads C. Thisstatement is a guide to understand the block-diagram depicted in FIG. 7.The input data stream D is converted into two streams D and itscomplement D by splitting and passing input data through a D-cell and aninverter (N). These data are then mixed in I and Q channels, which areeach formed by a pair of RSN cells. Set (S) and reset (R) inputterminals of each RSN pair are controlled by an LO code in such a waythat either D or its complement D data stream can go through the RSNs.There is no signal collisions at the mixer's I and Q outputs (MI and MQin FIG. 7) because either D or its complement D exist, and only one RSNcell of a pair is set by the applied code. The LO code in thisparticular 1×1-bit mixer implementation is generated by applying an LOclock (LOCLK) to a binary tree of toggle-flip-flops (T). The bit streamsshown next to the T cells illustrate how an input bit sequence fromLOCLK generates the LO codes C and C. Note also that the data clock CLKand LO clock LOCLK are distinct, and can be at different frequencies;the mixer elements themselves are asynchronous.

The mapping convention for the 1×1-bit mixer can be generalized to a1×k-bit mixer as illustrated in FIG. 3B. One can see that themultiplication function can be replaced by an bit-parallel exclusive-OR(XOR) operation and the mixing function can be rephrased as a simplerule: D ( D) reads out C (C).

The key element of a bit-slice of the 1×k-bit mixer (FIG. 5) is a pairof RSN cells with their states controlled by an LO code and read out byD or its complement D. The multi-bit mixer LO code can be quitecomplicated, and simple approaches like the binary tree of T-flip-flopsdo not work. To generate LO code for each slice, a shift-register (SR)based structure depicted in FIG. 8 was developed, which represents theinternal structure of the cell labeled SR at the right of FIG. 5. The SRis initialized by a set-local-oscillator (SLO) pulse sent to all SRstages. The resulting state of a particular D-cell depends on itsconnectivity with the SLO distribution line. The SLO pulse either goesin or passes by the D-cell of a particular SR stage. Then, the LO clock(LOCLK) reads out the content of the leftmost D-cell and shifts the restfrom right to left. Finally, all SR stages will be read out by LOCLKpulses and then be rewritten by another SLO pulse. Note that SLO pulsescan be derived from the LOCLK sequence by sending them to a frequencydivider (LO FD in FIG. 4). The required SLO pulse frequency depends onthe length of LO code representation. The propagation of SLO pulsesbetween bit-slices is controlled by the LOCLK as shown in FIG. 5. Notethat the LO sequence in FIG. 7 is hard-wired by the structure of thecells; in FIG. 8, the sequence is 1100, from left to right.

As shown in FIG. 5, the generated LO code goes to a pair of cellscomposed of a D-cell and an inverter N to be converted intocomplementary streams of C and C. States of RSN cells are controlled byC and C but read out by D and complement D as a result of performing amixing function.

FIG. 4 shows the block diagram of an entire I&Q mixer consisting of Iand Q channels and a synchronous pulse distribution network (SPDN) thatdelivers all pulses (D, D, LOCLK, SLO) to the I- and Q-mixers underclock control.

The 1×k-bit mixer described above can be converted into an n×1-bit mixerwithout making many changes to the building blocks. Assuming that ann-bit input data stream can be delivered to the I- and Q-mixers througha properly modified SPDN, and then convert each data bit into acomplementary pair of D and D inside a bit-slice, the block diagramdepicted in FIGS. 9 and 10 is arrived at. This differs from the 1×k-bitmixer only by the addition of a pair of D cell and an inverter on theleft side of block-diagram in FIG. 10.

The block diagram in FIG. 9 of the entire I&Q n×1-mixer looks similar to1×k-mixer, but the SPDN for the n×1-mixer distributes n-bit data insteadof a one-bit number and its complement. All pulses inside the SPDN arecontrolled by the clock and their delays to I- and Q-mixers areequalized.

Several chips were designed to test our k-bit and n-bit mixers and theircomponents. All chips were fabricated using the HYPRES superconductingNb Josephson junction IC process with critical current density J_(c)=4.5kA/cm². The chip layout of a 1×3-mixer is shown in FIG. 11. This matchesthe block-diagram in FIG. 4 except for the addition of frequency dividerCLKFD. This is used to generate the Nyquist clock and other pulsesrequired for the digital decimation filters (DDFs) that follow the I&Qmixer. They are delivered to the DDFs by the SPDN. The LO and CLKfrequency dividers were kept separate to avoid common locking of bothfrequencies.

For testing purposes a 3-bit LO code (I and Q components) with a lengthof 8 was used, as presented in FIG. 12A, with the time-dependence of theweighted magnitude of the 3-bit LO code plotted in FIG. 13A. Duringtesting, the LO code of I and Q LSBs was deliverately kept “all 1s” and“all 0s” and codes of the other two bits shifted by 90°. This artificialcode is read out to the RSN-pair-based mixing units (FIG. 4), thence tobe read out by either D or its complement D. If all applied data D are1s, the mixer's output is inverted code. If data are not applied (all0s), the mixer's output is just a replication of the stored LO code.

This superconducting 1×3-mixer chip was tested at a temperature T=4.2 K,using a low frequency (˜0.5 MHz) for convenience. FIG. 14 shows theresults, illustrating the correct behavior. Note that the correct outputsequence of 8 pulses appears after the SLO pulse with the predetermineddelay of 3 CLK pulses that is a signature of the 1×k-mixer's design.

A 3×1-mixer with an LO code (I and Q) as shown in FIG. 12B was designedand tested, with the time-dependence of the 1-bit LO code shown in FIG.13B. Note that the LO code is identical for all I or Q bits, but shiftedby 90° from each other. An automated low-frequency testing setup wasused to test this chip at low frequency (˜kHz). FIG. 15 illustrates thecorrect operation of the chip when a test sequence of four 3-bit “0s”and eight 3-bit “1s” is applied. Four “0s” are read out as LO code once,but eight “1s” are read out as inverted code two times in full agreementwith FIG. 12B.

Although the functionality of these chips was tested for convenience atlow frequencies, the circuit designs are expected to function up tofrequencies of order 20 GHz and higher.

Multi-bit mixers are parts of future digital radio frequency receiversthat also comprise ADCs and digital decimation filters with appropriateoutput drivers. FIG. 16 shows the layout of a LP ADR (all-digitalreceiver) on a 10×10 mm superconducting chip, with the mixer layout ofFIG. 11 located between the ADC on the left and the DDF on the right.

Multi-bit mixers of both types provide binary-weighted outputs, so theyshould be fed into the appropriate slices of the digital decimationfilter. The geometry of the ADR chip is configured in such a way thatthe l-bit output of I(Q)-mixer matches the l-bit input of thecorresponding I(Q)-DDF. Such an interface between the I(Q) mixer and acorresponding DDF also requires an additional adaptor block (the blocklabeled M2DDF in FIG. 16) to synchronize their timing. The DDF usedfeatures a clock skew of one clock per bit-slice. The interface isdesigned to delay the l-bit output of the mixer by sending it through aco-flow shift register with the number of stages equal to l. As aresult, the mixer's outputs are properly delayed and matched to thetiming of the DDF.

Multi-bit mixers provide an increased number of input bits for the DDF.That requires an increase in the number of bit-slices for the DDF. Thechip shown in FIG. 16 has two I and Q filters, each comprising 17bit-slices, that occupy almost all the chip space available for filtersin this particular configuration.

The embodiments of the multi-bit digital mixer presented aboveincorporate an integrated hard-wired digital LO circuit. These arecompact and efficient, but in some cases, a reprogrammable digital LOmay be desirable. This may be achieved using an external digitalgenerator, but the ultra-high-speed operation of this digital mixercircuit requires careful consideration of synchronization. In theapproach illustrated in FIG. 6, a large capacity room-temperaturesemiconductor memory was combined with a fast cryogenic RSFQ cache,integrated on the same chip with an RSFQ digital signal processor. Theasymmetric nature of the required memory operation—fast readout,infrequent addressing, no writing functions—allows utilization ofpipeline loading in order to avoid latency issues.

FIG. 6 shows a hybrid memory configuration and its integration into aDigital radio frequency channelizing receiver (such as that in FIG. 1B).This consists of a room-temperature high-capacity memory capable of fastreadout and a cryogenic superconducting RSFQ cache capable of receivingserial data, deserializing, and re-synchronizing the multiple bits. Thiscache memory is integrated with the In-phase (I) and Quadrature (Q)sections of a multi-bit digital mixer very similar to that shown in FIG.4.

For multi-bit mixing or correlation, the room-temperature memory shouldideally supply multi-bit words at the sampling clock frequency (20-30GHz). However, it is impractical to send all bits in parallel due toinevitable inter-bit jitter during transmission over the rather longdistance between a room-temperature memory module and a cryogenic RFDSP. To avoid this problem, data was serially supplied and then on-chipdata deserialization and re-synchronization performed. In FIG. 6, theserialized coded LO data is labeled SC, and the serializer clock islabeled SCCLK.

A particular objective was to find a commercially-available memory unitwith a relatively deep storage capacity (at least 64 Mbit) and capableof providing a 30 Gbps single-bit output data. These requirements matchwell to those of Arbitrary Bit Sequence Generators (ABSGs) such as theSympuls BMG 30G-64M 30 GBit/s pattern generator (see, for example,www.sympuls-aachen.de/en/bmg.html). This generates programmable binarysequences operating with an external clock generator between 1 and 30Gbit/s, with up to 64 Mb of memory.

The main function of the cache circuit is to receive high-speed serialdata (SC) from the ABSG at room temperature, perform datasynchronization (find end-of-word), and deserialize data into paralleloutput words for the digital mixer. To facilitate synchronization, thelast bit in the data word was reserved as the end-of-word bit.

The deserializing cache memory modules are integrated with the multi-bitdigital mixer on the same chip. In order to facilitate the integration,the previous design of the 1×k multi-bit mixer (FIGS. 4 and 5) wasmodified. FIG. 17 shows the block diagram adapted for integration withthe cache memory circuits. This comprises two deserializer (DS) units,one each for the I and Q channels, with a clock control unit (CU) thatcontrols both of them. The various clock and data signals aresynchronized in a synchronous pulse distribution network (SPDN) verysimilar to that in FIG. 4.

The cell-based design of single-bit slices of the mixer blocks in FIG.17 is shown in FIG. 18, similar to that in FIG. 5. Specifically, abuffer (the rightmost DFF) was placed between the deserializer and themixer for synchronizing the master clock (CLK) of the mixer with the LOcode C. The LO code is stored in RSN cells for several sampling clock(CLK) cycles until the next parallelized LO code arrives. With thistiming design, the cache and the mixer can operate under differentindependent clocks (SCCLK and CLK). Thus, the room-temperature memorydoes not have to be synchronized with the RSFQ processor master clock.The LO code loaded to the mixer LO inputs can be used for multiplecycles of RF data.

FIG. 19 shows a block diagram of the cache circuit. The deserializerpart (DS) of the cache design is based on a shift-and-dump demultiplexerarchitecture (similar to that in S. B. Kaplan et al., IEEE Trans. Appl.Supercond., vol. 5, pp. 2853-2856, June 1995, expressly incorporatedherein by reference) and implemented using dual-port DFF cells derivedfrom B flip-flops (described in S. Polonsky, et al., IEEE Trans. Appl.Supercond., vol. 4, pp. 9-18, March 1994, expressly incorporated hereinby reference). The data synchronization is performed by a clockcontroller circuit (CU) consisting of a static frequency divider and asynchronization circuit. The serialized data is received by a high-speedDC-to-SFQ converter and applied to a deserializer (DS). In this example,the clock controller splits every 8 pulses of external high-speed clockSCCLK into 7 serial clock pulses and 1 parallel read-out clock pulsewhich destructively reads out the content of the deserializer. The lastbit (sync bit) of the readout word is fed back to the clock controllerto provide data synchronization. If the synchronization bit has thewrong value, the clock controller shifts the read-out clock by oneperiod, searching for the end-of-word symbol, thus automaticallyrecovering the lost synchronization.

In a prototype device demonstration, due to space limitations on thetest chip (5 mm×5 mm), it was decided to limit the LO word length to 3bits for I and Q components. Consequently, the 8-bit cache was dividedinto a 4-bit block I-DS and a 4-bit block Q-DS in order to accommodate 1bit for synchronization, 6 bits for the LO code payload (3-bit I and3-bit Q), and 1 bit for monitor.

FIG. 20 shows the layout of a section of the prototype chip withintegrated cache circuit and the 1×3 digital I&Q mixer fabricated usingthe HYPRES standard superconducting niobium 4.5 kA/cm² process. The7-bit cache circuit is divided using a set of microstrip linestraversing the synchronous pulse distribution network (SPDN) section.This chip was tested at cryogenic temperatures of 4.2 K, and verifiedhigh-speed operation of all major components including separate cachecircuits and digital mixer blocks.

This design is adequate for digital radio frequency channelizingreceivers, since the LO code is relatively short. The loading rate ofthe room temperature memory limits the maximum frequency of the LO—for a30 Gbps loading rate, a 3.75 GHz maximum LO frequency can be realized.For higher LO frequencies, multiple parallel cache modules would have tobe used in a pipelined fashion. For longer templates necessary for adigital radio frequency cross-correlation receiver, a longer on-chipcache would be required.

The present embodiments are considered in all respects to beillustrative and not restrictive, and all changes which come within themeaning and range of equivalency of the claims are therefore intended tobe embraced within. The invention may be embodied in other specificforms without departing from the spirit or essential characteristicsthereof. The disclosure shall be interpreted to encompass all of thevarious combinations and permutations of the elements, steps, and claimsdisclosed herein, to the extent consistent, and shall not be limited tospecific combinations as provided in the detailed embodiments.

1. A multi-bit digital mixer comprising at least one Josephson junction,configured in a circuit to receive a first digital input signal and asecond digital input signal and to generate a multi-bit parallel outputsignal representing the multiplication product of the first digitalinput signal and the second digital input signal.
 2. The multi-bitdigital mixer of claim 1, wherein the second digital input signal isreceived from a digital local oscillator, and is generated using a shiftregister.
 3. The multi-bit digital mixer of claim 1, wherein the firstdigital input signal is communicated over at least one matched pair ofcomplementary binary inputs.
 4. The multi-bit digital mixer of claim 1,wherein the circuit comprises a plurality of bit slices, having acorresponding bit slice for each parallel bit of the output signal. 5.The multi-bit digital mixer of claim 1, wherein the circuit comprises atleast one respective XOR-based mixer cell for each bit of the multi-bitparallel output signal.
 6. The multi-bit digital mixer of claim 1,wherein the first digital input is received from a shift register. 7.The multi-bit digital mixer of claim 1, wherein at least one of thefirst and second digital input signals comprises a multi-bit parallelsignal which incorporates a synchronous pulse distribution network(SPDN) configured to ensure proper multibit timing.
 8. The multi-bitdigital mixer of claim 1, wherein the first digital input signal and thesecond digital input signal have respectively different clock rates. 9.The multi-bit digital mixer of claim 8, wherein the circuit comprises atleast one asynchronous RSFQ mixer.
 10. The multi-bit digital mixer ofclaim 1, wherein the first digital input signal receives a signalderived from an analog to digital converter, and the second digitalinput signal receives a signal derived from a digital local oscillator.11. The multi-bit digital mixer of claim 10, wherein the first digitalinput signal comprises a multi-bit signal.
 12. The multi-bit digitalmixer of claim 1, wherein the second digital input signal is receivedfrom a shift register configured to acts as a code generator.
 13. Themulti-bit digital mixer of claim 12, wherein the analog to digitalconverter receives a modulated radio frequency signal, and the circuitis configured to mix the modulated radio frequency signal to generatethe multi-bit parallel output signal representing at least a differencefrequency of the modulated radio frequency signal and the digital localoscillator.
 14. The multi-bit digital mixer of claim 1, wherein thecircuit is configured to generate the multi-bit parallel output signalrepresenting the multiplication product of the first digital inputsignal and the second digital input signal substantially without a carryoperation.
 15. The multi-bit digital mixer of claim 1, wherein thecircuit comprises a semiconductor circuit, configured to generate atleast one of the first digital input signal and the second digital inputsignal, further comprising an interface circuit configured to presentinformation from the semiconductor circuit to the circuit comprising theat least one Josephson junction.
 16. The multi-bit digital mixer ofclaim 1, wherein the circuit comprises a semiconductor circuit, furthercomprising an interface circuit configured to present information fromthe circuit comprising at least one Josephson junction to thesemiconductor circuit.
 17. The multi-bit digital mixer of claim 1,wherein the circuit is configured as a digital down converter of adigital radio frequency receiver receiving analog radio frequencyinformation modulated on a carrier having a carrier frequency of atleast 250 MHz, comprising a superconducting analog to digital converterhaving a sampling rate of at least 1 GHz and a multi-bit output.
 18. Themulti-bit digital mixer of claim 1, wherein the circuit is configured toproduce an updated multi-bit parallel output signal at a rate of atleast 20 GHz.
 19. The multi-bit digital mixer of claim 1, wherein thecircuit is configured as a receiver for receiving quadrature phasemodulated signals, further comprising an analog to digital converterdigitizing a radio frequency signal at a digital sample rate of at least1.5 GHz, wherein at least two local oscillator signals are provided byat least one shift register configured as a code generator to each of atleast two multi-bit digital mixers, each mixer being configured toproduce a multi-bit parallel output signal representing themultiplication product of the digital output of the analog to digitalconverter and a respective representation of the local oscillatorsignals, wherein the local oscillator signals are respectivelytime-shifted replicas of each other.
 20. A digital mixing method,comprising: receiving two asynchronous rapid single quantum flux (RSFQ)signals, at least one of the received RSFQ signals comprising aplurality of parallel bits, wherein each signal bit is represented ascomplementary signal pairs; ensuring proper timing of the plurality ofparallel bits of the two asynchronous RSFQ signals with a synchronouspulse distribution network (SPDN); and mixing the properly timed twoasynchronous RSFQ signals with an array of XOR-based asynchronous mixercells, at least one for each parallel bit line.